Method and system for synchronization in communication system

ABSTRACT

A method can be used for facilitating an uplink synchronization between a first transceiver and a second transceiver within a cell in a multi-user cellular communication system. The first transceiver receives a reference from the second transceiver and generates a set of signature sequences based on the reference. The first transceiver selects a first signature sequence from the set of signature sequences and incorporates the first signature sequence into a signal. The first transceiver transmits the signal to the second transceiver. The signal is used for a uplink synchronization between the first transceiver and the second transceiver. The set of signature sequences are generated from sequences with zero-correlation zone.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/175,685, entitled filed on Jul. 18, 2008, which is a continuation ofInternational Application No. PCT/CN2006/000077, filed on Jan. 18, 2006,both of which are incorporated herein by reference.

This application is also related to U.S. patent application Ser. No.13/560,877, which is filed concurrently herewith and is a continuationof the same priority chain.

TECHNICAL FIELD

The disclosure relates to the field of radio communication systems, andin particular to a method for uplink synchronization of a base stationand a mobile terminal in a multi-user cellular communication system.

BACKGROUND

In most mobile communication systems of today, there are specificrequirements regarding synchronization of a base station and a mobileterminal in order to secure a correct data transmission. Examples ofsuch systems are the Universal Terrestrial Radio Access (UTRA) andEvolved UTRA.

In Evolved UTRA, Single-Carrier Frequency Division Multiple Access(SC-FDMA) may be used as multiple access scheme for the uplinkcommunication. The transmission scheme of SC-FDMA is the so-calledDiscrete Fourier Transform-spread Orthogonal Frequency DomainMultiplexing (DFT-spread OFDM), which can be seen as OFDM withpre-coding. Whereas OFDM, which produces a multi-carrier signal, has ahigh peak-to-average ratio (PAPR), the DFT pre-coding gives asingle-carrier signal with lower PAPR. The low PAPR serves to extend thecoverage and to reduce the battery drain in the mobile.

In DFT-spread OFDM, cyclic prefix is used to achieve equalization in thefrequency domain. However, a requirement for successful equalization inDFT-spread OFDM, as well as in OFDM, is that the signals transmittedfrom all mobile terminals in the cell are synchronized in such a mannerthat the delay spread of the signal plus the spread in the time ofarrival is less than the duration of the cyclic prefix. Therefore, it isrequired that each transmitting mobile terminal is synchronized towithin a fraction of the duration of the cyclic prefix before it cantransmit data.

In Evolved UTRA, synchronization is performed both in uplink anddownlink. In one step of the synchronization, downlink synchronization,the mobile terminal synchronizes (or locks) to the carrier frequency andthe frame timing of the base station. This synchronization, however, isnot sufficient to ensure that the base station can properly receive thesignals from the mobile terminal, since mobile terminals may be locatedat various distances relative to the base station.

Consequently, further synchronization, uplink synchronization, is neededsince the distance between a base station and a mobile terminal, andhence the round trip time, is in general unknown.

In Evolved UTRA, a random access channel (RACH) supports uplinksynchronization of the mobile terminals. RACH in Evolved UTRA iscontention-based, i.e. any mobile terminal within the cell may transmiton the resource allocated to RACH. Consequently, several mobileterminals may attempt to transmit synchronization signalssimultaneously, and in order to reduce the risk that the base stationfails to distinguish signals from different mobile terminals, a set ofsignature sequences is provided, wherein each mobile terminal randomlyselects one signature sequence.

In UTRA and Evolved UTRA a binary pseudo-random sequence generated by ashift register is modulated by 16-bit Hadamard sequences to producethese signature sequences. Even though these signature sequences in manyinstances provide good correlation properties, there still exists a needfor enhanced detection capabilities to detect a specific signature inpresence of other simultaneous signatures, especially at low SIR values.

SUMMARY OF THE INVENTION

In one aspect, the disclosure provides a method and a system for uplinksynchronization in a multi-user cellular communication system, which hasenhanced capabilities to detect a single signature in presence of anumber of other simultaneous signatures, especially at lowsignal-to-interference ratio (SIR) values, as compared to the knownprior art.

In accordance with the present disclosure, a signature sequence istransmitted from a second transceiver to a first transceiver. Thesignature sequence is selected from a first set of signature sequences,and the received signal is correlated with at least one signaturesequence in the first transceiver to estimate the time of arrival tosynchronize transmission between the second transceiver and the firsttransceiver. The disclosure is characterized by a signature sequencecomprising, at least in part, a zero-correlation zone sequence. Thesignature sequence may be selected from a group of signature sequences.

The disclosure provides the advantage that, apart for maintainingfavorable features of prior art signature sequences, such as goodautocorrelation properties for allowing accurate timing estimation, goodcross-correlation properties to allow for accurate timing estimation ofdifferent simultaneous and partially synchronized signature sequences,and a small peak-to average power ratio, zero, or substantially zerocross-correlation for synchronous and simultaneous signature sequencesis achieved, which substantially improves the detection probability of aparticular signature sequence since the sequences are easilydistinguished from each other. The improved detection capabilitiesprovide the further advantage that in situations with more than onesimultaneously transmitted signature sequence, less retransmissions haveto occur due to missed detections, and, accordingly, system resourcesare more efficiently used. Further, as it is becoming more and moreimportant to obtain a fast access to the network and to be able toquickly transmit data using high power, the improved detectioncapabilities allow faster detection of a specific mobile terminal thatwishes to transmit data, which also facilitates interoperability withthe IP protocol.

The use of signature sequences according to the present disclosurefurther has the advantage that, even if the signal level of onesignature sequence is strong while the signal level of a substantiallysimultaneous signature sequence is considerably weaker, e.g., due todistance, shadowing or (perhaps most probably) fast fading, theprobability of a correct detection is substantially improved.

The zero-correlation zone of said first signature sequence may be of alength such that it substantially corresponds to the maximum expecteddelay of a transmission from the second transceiver to the firsttransceiver. Further, the received signal may be correlated with atleast one signature sequence for a predetermined number of delays of thesignal, e.g., corresponding to the maximum expected delay. The delay maybe determined using the cell size. This approach has the advantage thata desired length of the zero-correlation zone may be obtained, wherebythe number of signature sequences may be varied to provide the requiredzero-correlation zone length. The more sequences, the shorterzero-correlation zone.

A set of matched filters may be used in the first transceiver tocorrelate the received signal with at least one signature sequence oreach signature sequence in a group of signature sequences for apredetermined number of delays of the signals, whereupon a peak outputfrom each matched filter is detected, and after which the detected peakoutput from each filter is used to estimate the time of arrival tosynchronize the transmission from the second transceiver. This has theadvantage that the correlation may be performed in a simple manner.

The signature sequences may be taken from a set of GeneralizedChirp-Like sequences obtained by modulating a Zadoff-Chu sequence withan orthogonal set of complex sequences. For example, the orthogonal setof modulating sequences is a set of rows and/or columns of a discreteFourier transform matrix, or a set of rows and/or columns in a Hadamardmatrix. This has the advantage that the signature sequences may beaccomplished in a simple manner.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will become more readily apparent from the DetailedDescription of Illustrative Embodiments, which proceeds with referenceto the drawings, in which:

FIG. 1 shows a conventional DFT-spread OFDM transmitter structure forsynchronized data transmission;

FIG. 2 shows absolute values of the autocorrelation andcross-correlation functions of exemplary signature sequences accordingto the present disclosure;

FIG. 3 shows a magnified portion of the graph in FIG. 2 in greaterdetail;

FIG. 4 shows the probability of missed detection for one transmittedsequence according to the present disclosure; and

FIG. 5 shows the probability of missed detection for a transmittedsequence in the presence of one or more other transmitted sequencesaccording to the present disclosure.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The present disclosure will now be described more in detail in relationto a communication system employing DFT-spread OFDM.

In FIG. 1 is shown a basic transmitter 10 for DFT-spread OFDM. Blocks ofM complex modulated symbols x_(n), n=0,1, . . . ,M−1, are transformed bya DFT 11 which results in M coefficients X_(k):

$\begin{matrix}{{X_{k} = {\sum\limits_{n = 0}^{M - 1}{x_{n}{\mathbb{e}}^{{- {j2\pi}}\frac{nk}{M}}}}},{k = 0},1,\ldots\mspace{14mu},{M - 1.}} & (1)\end{matrix}$

The output from the DFT is mapped by a sub-carrier mapping module 12 onequidistant sub-carriers l_(k)=l₀+kL, where l₀ is a frequency offset,and L is an integer larger than or equal to 1. All other inputs to theN-point Inverse Discrete Fourier Transform (IDFT) are set to zero.

The output of the IDFT 13, y_(n), is given by

$\begin{matrix}{{y_{n} = {\frac{1}{M}{\sum\limits_{k = 0}^{M - 1}{X_{k}{\mathbb{e}}^{{j2\pi}\frac{{nl}_{k}}{N}}}}}},{n = 0},1,\ldots\mspace{14mu},{N - 1}} & (2)\end{matrix}$

Finally, to avoid inter-symbol interference (ISI) and inter-channelinterference (ICI), a cyclic prefix inserter 14 inserts a cyclic prefix,i.e., a copy of the last portion of each OFDM symbol is inserted beforethe beginning of the same symbol. A time window may be applied after thecyclic prefix to reduce out-of-band emissions.

The cyclic prefix enables equalization in the frequency domain. However,a requirement for successful equalization in DFT-spread OFDM, as well asOFDM, is synchronization of the transmitted signals from all mobileterminals in a cell so that the delay spread of the signal plus thespread in the time of arrival is less than the duration of the cyclicprefix. It is therefore required that the mobile terminal issynchronized to within a fraction of the duration of the cyclic prefixbefore it can transmit data.

As stated above, in a first synchronization step in a DFT-spread OFDMsystem, the mobile terminal uses the carrier frequency and frame timingof the base station to perform synchronization. Although thissynchronization step ensures that the downlink synchronized mobile canreceive the signals from the base station, further synchronization isneeded to compensate for the, generally unknown, distance between themobile terminal and the base station so as to ensure that the basestation can properly receive the signals from the mobile terminal. Amobile terminal far away from the base station will receive downlinksignals with a larger delay than a mobile terminal close to the basestation and the transmitted signals in uplink will take longer time topropagate to the base station for this mobile terminal, as compared tosignals from a mobile terminal closer to the base station. Once the basestation has estimated the time it will take for a signal transmittedfrom the mobile terminal to reach the base station, the base station maytransmit a command to the mobile terminal to adjust its transmit timingso that transmissions from various mobile terminals arrive at the basestation at desired points in time.

An important aspect of the second step of synchronization is that themobile terminals have already synchronized the reception of the downlinksignal and that all variations in time of arrival at the base station ofthe signals transmitted from the mobile terminals are due to thedifferent round-trip times. Since the cell size is known, the range oftime of arrivals is known a priori in the base station.

In Enhanced UTRA, the random access channel (RACH) in uplink supportsuplink synchronization of mobile terminals. It is mapped onto certainresources in time (access slots) and frequency. In each access slotthere should be a guard interval, so that all the transmitted signalsarrive within the allocated time and do not interfere with datatransmissions no matter where the transmitting mobile terminal islocated in the cell. Since the RACH in Evolved UTRA is contention-based,i.e. any mobile terminal within the cell may transmit on thetime-frequency resource allocated to RACH, more than one mobile terminalmay simultaneously, or substantially simultaneously, attempt to transmitsynchronization signals. In order to reduce the risk that the basestation fails to distinguish the signals from the different mobileterminals, a set of signature sequences is used, wherein each mobileterminal, normally in a random fashion, selects one signature sequenceout of the set of signature sequences.

Since successful detection of the signature sequence is necessary forthe mobile terminal to access the network, it is important that thetransmitted signature sequence requires a low power amplifier back-offto allow for high average transmit power and hence good coverage.

The signature sequences in uplink should have the following properties:

good autocorrelation properties to allow for accurate timing estimation,

good cross-correlation properties to allow for accurate timingestimation of different simultaneous and partially synchronized (i.e.downlink synchronized) signature sequences, wherein the phase differenceis limited by the maximum round-trip time in the cell,

zero cross-correlation for synchronous and simultaneous signaturesequences, and

a small peak-to-average power ratio.

These properties are satisfied to a large extent by the RACH signaturesin UTRA used today, and, at least partially, these also constitute thecurrent suggestion for Evolved UTRA. In UTRA, a binary pseudo-randomsequence generated by a shift register is modulated by 16-bit Hadamardsequences to produce the signature sequences. Further, rotation of thesignal constellation is applied to reduce the PAPR of the signal.

The modulation using Hadamard sequences allows for complexity reductionin the receiver: For each delay, the received signal is multipliedelement-wise with the complex conjugate of the pseudo-random scramblingsequence. Every 16^(th) sample is summed to produce a vector of 16elements. Finally, the Hadamard sequences are correlated with thereceived vector to produce correlation outputs of the signaturesequences.

However, some of the properties of these known signature sequences, suchas mutual cross-correlation, i.e., the detection probability of a singlesignature in presence of one or more other simultaneous signatures couldbe better, especially at low SIR values.

According to the present disclosure, the above problem is overcome byusing zero-correlation zone sequences, i.e., a downlink synchronizedmobile terminal transmits a signal, which is a signature sequence from aset of zero-correlation zone sequences.

A set of M sequences {d_(x)(k)}, x=0 ,1, . . . , M−1, k=0,1, . . . ,N−1, of length N is said to be a set of zero-correlation zone sequencesif all the sequences in the set satisfy the following autocorrelationand cross-correlation properties:

The periodic autocorrelation function Σ_(k=0) ^(N−1)d_(x)(k)d*_(x)((k+p)mod N) is zero for all p such that 0<|p|≦T and the periodiccross-correlation function Σ_(k=0) ^(N−1)d_(x)(k)d*_(y)((k+p) mod N) iszero for all p such that |p|≦T (including p=0). T is the length of thezero-correlation zone.

In an exemplary embodiment of the invention, the set of zero correlationzone sequences is constructed by using Generalized Chirp-Like (GCL)sequences. A GCL sequence {c(k)} is defined asc(k)=a(k)b(k mod m), k=0,1, . . . , N−1.  (3)where N=sm², s and m are positive integers, {b(k)} is any sequence of mcomplex numbers of unit magnitude, and {a(k)} is the Zadoff-Chu sequence

$\begin{matrix}{{a(k)} = \left\{ {{{\begin{matrix}{W_{N}^{{k^{2}/2} + {qk}},} & {N\mspace{14mu}{even}} \\{W_{N}^{{{k{({k + 1})}}/2} + {qk}},} & {{N\mspace{14mu}{odd}},}\end{matrix}k} = 0},1,\ldots\mspace{14mu},{N\text{-}1},{q\mspace{14mu}{is}\mspace{14mu}{any}\mspace{14mu}{integer}},} \right.} & (4)\end{matrix}$where W_(N)=exp(−j2πr/N) and r is relatively prime to N (i.e., thegreatest common divisor of r and N equals 1).

Any GCL sequence has an ideal periodic autocorrelation function, i.e. itis a Constant Amplitude Zero Auto-Correlation (CAZAC) sequence.

If the two GCL sequences c_(x)(k) and c_(y)(k) are defined by using thesame Zadoff-Chu sequence {a(k)} but different, arbitrary modulationsequences {b_(x)(k)} and {b_(y)(k)}, it can be shown (in a mannersimilar to what is disclosed in B. M. Popovic, “New Complex Space-TimeBlock Codes for Efficient Transmit Diversity,” IEEE 6th Int. Symp. onSpread-Spectrum Tech. & Appl (ISSSTA 2000)., NJ, USA, pp. 132-136,September 2000.) that the periodic cross-correlation is zero for alltime shifts p in the delay zones0<|p|<sm, sm<|p|<2sm, . . . , (m−1)sm,<|p|<sm ².

Thus, if the above two modulation sequences are orthogonal, theresulting GCL sequences will be not just orthogonal, but also will havea zero-correlation zone of length sm−1.

Based on this property, the set of m zero correlation zone sequences canbe defined as the set of GCL sequences obtained by modulating a commonZadoff-Chu sequence {a(k)} with m different orthogonal modulationsequences {b_(i)(k)}, i=0,1,2, . . . , m−1, k=0,1,2, . . . , m−1. Theperiodic cross-correlation between any two sequences from the set willbe zero for all the delays between −sm and +sm.

The sequences from the set of zero-correlation zone sequences are usedas the synchronization signatures. Although the matched filters for suchsignatures actually calculate the aperiodic cross-correlations, it isexpected that the ideal periodic cross-correlation properties in thesearch window will be to a large extent preserved. The reason is thatfor delays in the search window that are much smaller than the length ofthe sequence, the sums for the aperiodic and periodic cross-correlationvalues only differ in a small number of terms. This expectation isconfirmed by numerical evaluations, as will be shown later.

For the GCL sequences, possible choices for the selection of orthogonalmodulation sequences would, for example, be either the sets of Hadamardsequences or Discrete Fourier Transform (DFT) sequences. The set of DFTsequences is defined asb _(i)(k)=W _(m) ^(ik) , i,k=0,1, . . . , m−1,  (5)while the set of Hadamard sequences is defined as the rows (or columnsor possibly both rows and columns) in a m×m Hadamard matrix, defined asfollows: A Hadamard matrix H_(m) of order m, consists of only 1s and −1sand has the property H_(m)H_(m) ^(T)=mI where I is the identity matrixand ^(T) denotes transpose. Hence, the Hadamard sequences areorthogonal. For m=2^(n), where n is a positive integer, Hadamardsequences can be defined as

$\begin{matrix}{{{b_{i}(k)} = \left( {- 1} \right)^{\sum\limits_{l = 0}^{m - 1}{i_{l} \cdot k_{l}}}},i,{k = 0},1,{{\ldots\mspace{14mu} m} - 1},} & (6)\end{matrix}$where i_(l), k_(l) are the bits of the m-bits long binaryrepresentations of integers i and k.

The actual numbers m and N can be selected to fit into the requirementsof the Evolved UTRA. For a given length of the sequence, there is then atradeoff between the length of the zero-correlation zone and the numberof signatures that can be provided.

For example, for a 1.25 MHz bandwidth in Evolved UTRA, an exemplary timeavailable for signature sequence transmission is 500 μs, and with aguard time of about 110 μs, the duration of the sequence is 390 μs.Assuming a sampling rate of, e.g., 1.024 MHz, it follows that the lengthof the sequences isN=400=sm².

The cell size is generally known, and thereby the maximum timedifference between signals from two mobile terminals in the cell (i.e.,the sum of the additional propagation times to and from the one mobileterminal relative to the other). Advantageously, the zero-correlationzone length is adapted to this time difference, i.e. to obtain a lowcorrelation for all possible time differences up to the maximum possibledifference. If, for example, the cell size is 14 km, the maximum traveltime for a signal corresponds to 96 symbols with the above presumptions.The low cross-correlation in this delay range will, according to theabove, be ensured if sm=100, so it follows that m=4, and s=25 (a largerm would result in a shorter, and thereby unsatisfactory,zero-correlation zone length). For simplicity, we choose q=0 in equation(4). It is to be understood, however, that other values of q may beused. A non-zero q will cause a shift in the sequence. Hence, there are4 different signature sequences of length 400.

In FIG. 2 is shown the absolute values of the autocorrelation andcross-correlation functions of the sequences with the received signal.

The amplitude of the aperiodic cross-correlation function

${{R_{xy}(p)} = {\sum\limits_{k = 0}^{N - 1 - p}{{c_{x}(k)}{c_{y}^{*}\left( {k + p} \right)}}}},$where p is the delay and “*” denotes complex conjugate, is shown in FIG.2 for the DFT-modulated GCL sequence, with N=400 (s=25 and m=4), andr=1.

A set of Hadamard-modulated GCL sequences has autocorrelation andcross-correlation functions similar to the ones shown in FIG. 2. Thepeaks of the cross-correlation functions are located near multiples ofsm=100. The peaks exhibit a certain broadening, i.e. the correlationvalues close to multiples of sm have considerable non-zero values, whichare not otherwise present for the periodic cross-correlation functions.However, for the given parameters, the cross-correlation functions donot exceed 20 for delays less than 96. Accordingly, for a cell of size14 km, only the portion of the plot up to p=96 is of interest, and inthis interval the result of the correlation is unambiguous. The portionof the plot in FIG. 2 showing delays from 0 to 100 is shown more indetail in FIG. 3.

The actual synchronization is performed by the base station using a setof matched filters to correlate the received signal with the signaturesequences in the set of signature sequences for all delays within thesearch window, and detecting a peak output from each matched filter. Athreshold is used to reduce the probability of false detection, i.e.,the threshold is set to a value such that when the received signal onlyconsists of noise it results in a detection with a certain probability,e.g. 0.0001.

The detected peak output from each filter is then used to estimate thetime of arrival, i.e. the delay, to synchronize the transmission fromthe mobile terminal.

The comparison signal in the base station may be non-periodic, i.e.,consist of only one period. Alternatively, this signal may be periodicor consist of one period plus a portion of a period on either or bothsides. If a periodic signal is used, the threshold must be increasedsince the probability of an erroneous detection increases. On the otherhand, the robustness is increased when more than one signature sequenceare present. Further, it is, of course, also possible to extend thesignature sequence transmitted by the mobile terminal a portion of aperiod on either or both sides of the sequence. The length of theadditional portion(s) may be determined by the time available fortransmitting the signature sequence.

In one embodiment of the present invention, all cells in a system areprovided with the same number of signature sequences, preferably thisnumber is selected based on the largest cell in the system. As isapparent, however, the specific signature sequences may vary from cellto cell. This has the advantage that when a mobile terminal is presentat the border between two cells, it can be determined which cell ittries to connect to. If neighboring cells have the same set of signaturesequences, two or more base stations may attempt to answer the call fromthe mobile terminal. On the other hand, it may be determined which basestation provides the best signal quality, and thereby which base stationshould answer. As also is apparent from the above, however, it is alsopossible to have different sets of signature sequences in differentcells. The various sets of signature sequences can easily be obtained byvarying r. Which r value to use may be transmitted to the mobileterminal, which thereby can produce the set of signature sequencesaccording to the equations above. Further, if the cell size is smaller,the number of signature sequences may be increased with maintainedsequence length. If, for example, the cell size is 7 km, the number ofdelay steps needed is only half of the above example. Accordingly, m canbe set to 7 and s to 8. This will result in a signature sequence oflength 392, and 7 signature sequences fulfilling the requirements of thecell size, i.e. during sm−1=55 steps. In this example, the above guardtime has been maintained. It is, however, also possible to reduce theguard time in smaller cells and thereby enable longer signaturesequences, and consequently also an increased number of sequences.

The detection performances of these proposed signature sequences, orpreambles, have been evaluated by link-level simulations. The truncatedWCDMA RACH preamble has been used as a reference with modulatingHadamard sequences that are 4 bits long, instead of 16 bit longsequences, to keep the same number of signature sequences as for theproposed sequences. The number of receive antennas is two andcorrelations from the two antennas at the same delay are combinednon-coherently, i.e., the absolute values of the squared matched filteroutputs from the two antennas at the same delay are added. The number oftrials is 100000.

Two scenarios have been simulated. In both scenarios the detectorcorrelates the received signal with all possible signature sequences inthe search window. A threshold is set to give a false alarm probabilityof 0.0001 for a signature sequence at a single delay. Missed detectionis declared if the transmitted signature sequence is not detected.

In the first scenario, only one preamble is transmitted in atime-frequency resource for RACH. The delay is randomly distributedwithin the search window, i.e., in this example, ranging from 0 to 96samples, corresponding to randomly distributed mobiles in the cell.

In the second scenario, two or more different signature sequences fromthe same set are transmitted in the same time-frequency resource. Thesignal-to-noise ratio (SNR) of signature S1 is fixed (SNR=−15 dB) andthe other interfering signatures are transmitted with various poweroffsets to signature 1. However, all interfering signatures aretransmitted with the same power. All signatures are transmitted withindependent random delays within the search window. The probability ofmissed detection of the weaker signal, signature S1, is recorded. TheSIR is the ratio of the power of signature S1 to the power of any of theinterfering signatures.

Simulation results are shown for scenarios 1 and 2 in FIGS. 4 and 5,respectively. In FIG. 4, the probability of missed detection for onetransmitted sequence is shown, and in FIG. 5 the probability of misseddetection of a transmitted sequence in presence of another transmittedsequence is shown. From FIG. 4 it is clear that there is no differencein the probability of missed detection in the case without aninterfering sequence as compared to the prior art. Hence, in thissituation, the signature sequences according to the present disclosureperform as well as the prior art sequences.

Regarding the second scenario, however, with two or more simultaneously,or substantially simultaneously transmitted sequences, the results shownin FIG. 5 clearly demonstrate significantly improved detectionperformance in the presence of one or several interfering sequences forthe set of sequences according to the present disclosure. For theproposed set of sequences, the detection performance does not changewith an increased number of interferers, not even for very low SIRvalues, whereas for the reference sequences, the performancedeteriorates substantially, both as the number of interferers increases,and with decreasing SIR. This substantial difference can, at leastpartially, be explained by the condition that when a strong signal and aweak signal are simultaneously present, parts of the stronger signalwill, during correlation, be interpreted as part of the weaker signal,with an incorrectly calculated delay as result. The use of signaturesequences according to the present disclosure has the advantage that, ascan be seen in FIG. 5, even if the signal level of one signaturesequence is strong while the signal level of a substantiallysimultaneous signature sequence is considerably weaker, the probabilityof a correct detection is substantially improved.

The lower probability of missed detection exhibited for the proposed setof sequences is due to the good cross-correlation properties of thezero-correlation zone sequences, and consequently, the presentdisclosure provides a substantial improvement as compared to the priorart. Further, this improvement of the detection probability by the useof zero-correlation zone sequences can allow reduction of thetransmitted power for RACH preamble, thereby reducing the overallinterference in the system and increased battery life.

Further, in the above description the disclosure has been described asutilizing full zero-correlation zone sequences. It is, however, alsopossible to use truncated sequences, i.e, not all of thezero-correlation zone sequences are used. This will reduce the detectionprobability, however with the advantage that the freedom in selectingnumber of signature sequences for a particular signature lengthincreases. The truncation may vary with the cell size. In smaller cellsa larger truncation may be accepted with maintained satisfactoryperformance.

As has been disclosed above, the present disclosure has severaladvantages. There are, however, other characteristics that have to beconsidered in order for the system to operate properly. For example, ashas been mentioned above, it is important that the transmitted signaturesequence requires a low power amplifier back-off to allow for highaverage transmit power and hence good coverage. Two measures related tothe power back-off are the peak-to-average power ratio (PAPR) and thecubic metric (CM).

In the following, the impact of the present disclosure on these measureswill be disclosed.

Let z(t) be the normalized baseband signal, such that its expectationvalue E(Iz(t)|²)=1. The PAPR at the 99.9^(th) percentile is defined asthe value x such that the probability that 10 log₁₀(Iz(t)|²)<×equals0.999.

The CM is defined asCM=[20 log₁₀((v_norm³)_(rms))−20 log₁₀((v_norm_ref³)_(rms))]/1.85  (7)where

v_norm is the normalized voltage waveform of the input signal

v_norm_ref is the normalized voltage waveform of the reference signal(12.2 kbps AMR Speech in WCDMA).

Table 1 lists the PAPR values at the 99.9^(th) percentile for areference WCDMA RACH preamble truncated to 400 samples with 4-bitHadamard modulating sequences, and for the GCL sequences with DFT andHadamard modulating sequences. Table 2 lists the corresponding CMvalues.

TABLE 1 PAPR (99.9th percentile) values Pulse-shaping filter WCDMAGCL-DFT GCL-Hadamard Sinc 3.9-5.9 dB 2.8 dB 4.5 dB Root-raised cosine,2.6-3.4 dB 3.0 dB 3.6 dB roll-off factor = 0.15

TABLE 2 Cubic metric values Pulse-shaping filter WCDMA GCL-DFTGCL-Hadamard Sinc 0.1-0.5 dB −0.6 dB 1.4 dB Root-raised cosine, −0.3 to−0.1 dB −0.6 dB 1.1 dB roll-off factor = 0.15

In all cases the maximum PAPR value is given over all modulatingsequences. The range of values given for the WCDMA RACH preamble is overall scrambling codes. For the GCL sequences, the Zadoff-Chu sequencewith r=1 has been used. It is to be understood that this specificexample of the value of r only is exemplary. Two different pulse-shapingfilters are applied, a simple sinc filter and a root-raised cosinefilter with roll-off factor 0.15.

From the tables, it is clear that the DFT-modulated sequence has bothlower PAPR and lower cubic metric than the Hadamard-modulated GCLsequence. Furthermore, applying a root-raised cosine filter improvesneither PAPR nor the cubic metric of the DFT-modulated sequence.

Finally, the PAPR of the DFT-modulated GCL sequence is essentially asgood as the WCDMA sequences with a root-raised cosine filter, while thecubic metric is somewhat better than for the WCDMA sequences.Apparently, it is possible to find sets of zero-correlation zonesequences that allow for a low power back-off.

What is claimed is:
 1. A method for facilitating an uplinksynchronization between a first transceiver and a second transceiverwithin a cell in a multi-user cellular communication system, the methodcomprising: receiving, by the first transceiver, a reference from thesecond transceiver; generating, by the first transceiver, a set ofsignature sequences based on the reference; selecting, by the firsttransceiver, a first signature sequence from the set of signaturesequences; incorporating, by the first transceiver, the first signaturesequence into a signal; and transmitting, by the first transceiver, thesignal to the second transceiver, wherein the signal is used for auplink synchronization between the first transceiver and the secondtransceiver, the set of signature sequences being generated fromsequences with zero-correlation zone, and the sequences withzero-correlation zone being obtained from a Zadoff-Chu sequence, theZadoff-Chu sequence being: ${a(k)} = \left\{ {\begin{matrix}W_{N}^{{k^{2}/2} + {qk}} & {,{N\mspace{14mu}{even}}} \\W_{N}^{{{k{({k + 1})}}/2} + {qk}} & {,{N\mspace{14mu}{odd}}}\end{matrix},{k = 0},1,\ldots\mspace{14mu},{N - 1},{{{where}W_{N}} = {\exp\left( {{- j}\; 2\pi\;{r/N}} \right)}},} \right.$where r, q and N are integers and r is relatively prime to N.
 2. Themethod of claim 1, wherein each of the sequences with zero-correlationzone is obtained by modulating the Zadoff-Chu sequence with anorthogonal modulation sequence.
 3. The method of claim 1, wherein thereference received from the second transceiver is designated for theparameter r.
 4. The method of claim 1, further comprising incorporatinga cyclic prefix into the signal.
 5. The method of claim 1, wherein thefirst signature sequence is used in a preamble portion of the signal andthe signal is transmitted over a random access channel.
 6. The method ofclaim 1, further comprising receiving a command from the secondtransceiver, wherein the command corresponds to a time of arrival of thesignal from the first transceiver to the second transceiver and thecommand is used for adjusting a transmission timing of the firsttransceiver.
 7. The method of claim 2, wherein the orthogonal modulationsequence is a Hadamard sequence originated from a Hadamard matrix Hm oforder m, the Hadamard matrix consists of only 1 and/or −1, and theHadamard matrix has the property HmHmT=mI, where I is the identitymatrix, m is an integer and T denotes transpose.
 8. An apparatusoperable to communicate in a wireless communications system, theapparatus comprising a processor coupled to a memory, wherein theprocessor is programmed to operate in the wireless communication systemby: receiving a reference from a transceiver; generating a set ofsignature sequences based on the reference; selecting a first signaturesequence from the set of signature sequences; incorporating the firstsignature sequence into a signal; and transmitting the signal to thetransceiver, wherein the signal is used for a uplink synchronizationbetween the apparatus and the transceiver, the set of signaturesequences being generated from sequences with zero-correlation zone, andthe sequences with zero-correlation zone being obtained from aZadoff-Chu sequence, the Zadoff-Chu sequence being:${a(k)} = \left\{ {\begin{matrix}W_{N}^{{k^{2}/2} + {qk}} & {,{N\mspace{14mu}{even}}} \\W_{N}^{{{k{({k + 1})}}/2} + {qk}} & {,{N\mspace{14mu}{odd}}}\end{matrix},{k = 0},1,\ldots\mspace{14mu},{N - 1},{{{where}W_{N}} = {\exp\left( {{- j}\; 2\pi\;{r/N}} \right)}},} \right.$where r, q and N are integers and r is relatively prime to N.
 9. Theapparatus of claim 8, wherein each of the sequences withzero-correlation zone is obtained by modulating the Zadoff-Chu sequencewith an orthogonal modulation sequence.
 10. The apparatus of claim 8,wherein the reference received from the transceiver is designated forthe parameter r.
 11. The apparatus of claim 8, wherein the processor isfurther programmed to incorporate a cyclic prefix into the signal. 12.The apparatus of claim 8, wherein the first signature sequence is usedin a preamble portion of the signal, and the signal is transmitted overa random access channel.
 13. The apparatus of claim 8, wherein theprocessor is further programmed to receive a command from thetransceiver, wherein the command corresponds to a time of arrival of thesignal from the apparatus to the transceiver and wherein the command isused for adjusting a transmission timing of the apparatus.
 14. Theapparatus of claim 9, wherein the orthogonal modulation sequence is aHadamard sequence originated from a Hadamard matrix H_(m) of order m,the Hadamard matrix consisting of only 1 and/or −1, and the Hadamardmatrix having the property H_(m)H_(m) ^(T)=mI, where I is the identitymatrix, m is an integer and T denotes transpose.
 15. A wirelesscommunication system, comprising: a first transceiver; and a secondtransceiver, wherein the first transceiver comprises a processor coupledto a memory, wherein the processor is programmed to: receive a referencefrom the second transceiver; generate a set of signature sequences basedon the reference; select a first signature sequence from the set ofsignature sequences; incorporate the first signature sequence into asignal; and transmit the signal to the second transceiver, wherein thesignal is used for a uplink synchronization between the firsttransceiver and the second transceiver, the set of signature sequencesbeing generated from sequences with zero-correlation zone, and thesequences with zero-correlation zone being obtained from a Zadoff-Chusequence, the Zadoff-Chu sequence being:${a(k)} = \left\{ {\begin{matrix}W_{N}^{{k^{2}/2} + {qk}} & {,{N\mspace{14mu}{even}}} \\W_{N}^{{{k{({k + 1})}}/2} + {qk}} & {,{N\mspace{14mu}{odd}}}\end{matrix},{k = 0},1,\ldots\mspace{14mu},{N - 1},{{{where}W_{N}} = {\exp\left( {{- j}\; 2\pi\;{r/N}} \right)}},} \right.$where r, q and N are integers and r is relatively prime to N.
 16. Thewireless communication system of claim 15, wherein each of the sequenceswith zero-correlation zone is obtained by modulating the Zadoff-Chusequence with an orthogonal modulation sequence.
 17. The wirelesscommunication system of claim 15, wherein the reference received fromthe second transceiver is designated for the parameter r.
 18. Thewireless communication system of claim 15, wherein the first signaturesequence is used in a preamble portion of the signal, and the signal istransmitted over a random access channel.
 19. The wireless communicationsystem of claim 15, wherein the processor is further programmed toreceive a command from the second transceiver, the command correspondingto a time of arrival of the signal from the first transceiver to thesecond transceiver, and the command being used for adjusting atransmission timing of the first transceiver.
 20. The wirelesscommunication system of claim 16, wherein the orthogonal modulationsequence is a Hadamard sequence originated from a Hadamard matrix H_(m)of order m, the Hadamard matrix consists of only 1 and/or −1, and theHadamard matrix has the property H_(m)H_(m) ^(T)=mI, where I is theidentity matrix, m is an integer and T denotes transpose.